Compact Planar Dual-Wideband Bandstop Filters with Cross Coupling and Open-Ended Stepped Impedance Resonators

148 Vamsi Krishnaet al. © 2010 ETRI Journal, Volume 32, Number 1, February 2010

This letter presents the design of a compact bandstop filter (BSF) operating at two frequencies. The proposed BSF consists of open-ended stepped impedance resonators (OSIR) and an end-shorted parallel-coupled microstrip line (E-PCML). The OSIRs are used to achieve the impedance-controlled stopband positions. The wide BSF bandwidths are achieved through enhanced coupling of the E-PCML. Explicit design guidelines are derived using a lossless transmission line model. To validate theoretical predictions, a prototype dual-band BSF operating at 900 MHz and 2,100 MHz with fractional bandwidths of 72% and 36%, respectively, is implemented in microstrip.

Keywords: Microstrip filter, bandstop filter, dual-band, open- ended stepped impedance resonator, coupled line.

I. Introduction

Recent trends in multiband wireless communication system applications, such as global systems for mobile communications, wireless local area networks, and many wireless front-end applications, require compact dual-band bandstop filters (DBBSFs) for concurrent interference suppression at two frequencies. The DBBSF was presented in [1], applying frequency-variable transformations to a prototype low-pass filter.

Capacitor-loaded tunable resonators (stubs) in combination with parallel-coupled resonators were proposed to realize multiple narrow stopbands in [2]. Lumped-element composite shunt resonators were used in a DBBSF design proposed in [3], which

Manuscript received July 13, 2009; revised Oct. 8, 2009; accepted Oct. 19, 2009.

Vamsi Krishna Velidi (phone: +91 322 2281 460, email: vvamsi.iitkgp@gmail.com) and Subrara Sanyal (email: ssanyal@ece.iitkgp.ernet.in) are with the Indian Institute of Technology Kharagpur, West Bengal, India.

doi:10.4218/etrij.10.0209.0341

incorporates parallel-connected open stubs with different lengths and widths. In [4], a DBBSF was realized by replacing the parallel-connected quarter-wavelength (λ/4) open stubs in the conventional design by stepped impedance resonators (SIR).

In this work, a simple and compact DBBSF is proposed which uses a single end-shorted parallel-coupled microstrip line (E- PCML) and open-ended SIRs (OSIRs). The stopband positions are easily controlled by varying the impedance ratio of the OSIRs. The bandwidths are controlled by the coupling gap of the E-PCML. The design procedure is simple and straightforward.

II. Analysis and Design of the Filter

Figure 1 shows the configuration of the proposed DBBSF.

The filter design can be separated into two parts, namely, the E- PCML and the OSIR. At the midband frequency (fC), the E- PCML unit is λ/4 long and exhibits an all-pass response [5].

The even- and odd-mode characteristic impedances (electrical lengths) of the E-PCML are Z0e and Z0o (θe and θo), respectively.

For simplicity, equal electrical lengths (θe=θo=θ) are considered.

Double-section OSIRs (λ/2 long at fC) are added at the feed line connecting points of the E-PCML to obtain two stopbands at symmetrical distances from fC. The OSIR characteristic impedances (electrical lengths) are Z1 and Z2 (θ1 and θ2). Using a lossless transmission line model, the transmission coefficient of the filter is derived as

[ ] (

)

^{( )}

21 2 2

0e 0o 0 0 0e 0o

2 ,

2 2

Z Z Z

S = M NP j N Z Z Z PZ Z Z M NP

⎡ ⎤

+ + ⎣ + − + ⎦

(1) where M =

(

)

Compact Planar Dual-Wideband

Bandstop Filters with Cross Coupling and Open-Ended Stepped Impedance Resonators

Vamsi Krishna Velidi and Subrata Sanyal

ETRI Journal, Volume 32, Number 1, February 2010 Vamsi Krishnaet al. 149 Fig. 1. Configuration of the proposed DBBSF.

Z2, θ2

Z1, θ1

Z2, θ2

Z1, θ1

Z0e, θ Z0o

1 2

Fig. 2. Variation of stopband frequency ratio and positions with Z₂/Z₁.

f2/f1

0 1 2 3 4 5

0 1 2 3 4 5

Frequency ratio

ZR=Z2/Z1

fZ/fC

( )

0e 0o 0e 0o

2 cot tan ,

N= Z Z Z θ+Z θ

(

) (

)

The transmission zero positions specify the stopband positions and are obtained by setting |S21|=0 as

2 1tan tan .1 2

Z =Z θ θ (2) At fC, the electrical lengths are θ1=θ10 and θ2=θ20. The two stopbands are at symmetrical distances from fC when θ10=θ20= π/2. The resonant frequencies are given by

( ) ( )

1 1

1 C R 2 C R

2 2

tan , tan ,

f f Z f f π Z

π ⁻ π ⎡ ⁻ ⎤

= = ⎣ − ⎦ (3)

where fC =

(

)

Figure 2 shows the variation of stopband positions fZ/fC

versus the OSIR impedance ratio ZR (dotted line). The zero separation decreases with increasing ZR. Figure 2 also shows the ratio f2/f1 versus ZR when θ1=θ2 (solid line). The impedance ratio ZR is determined by the desired stopband frequency ratio.

For the mobile bands centered at 0.9 GHz and 2.1 GHz, f2/f1=2.33, fC=1.5 GHz, and, from Fig. 2, ZR=1.94. The advantage of the present configuration is that the input signal transfers to the output not only through the λ/4 long parallel lines of the E-PCML, but also through the mutual coupling between these two closely spaced lines. Smaller spacing

Fig. 3. Computed responses of DBBSF (Z_R=2).

0 0.4 0.8 1.2 1.6 2.0 2.4

-50 -40 -30 -20 -10 0

f/f0

Z0e = 60 Ω, Z0o = 30 Ω

|S21| (dB)

Z0e = 100 Ω, Z0o = 50 Ω Z0e = 140 Ω, Z0o = 70 Ω

Fig. 4.Fractional bandwidth variation in the first stopband with impedances of E-PCML (ZR = 2).

50 100 150 200

50 60 70 80 90

% 3 dB FBW Z0o=90 Ω

Z0o=70 Ω Z0o=50 Ω

Z0o (Ω)

between the coupled lines results in stronger coupling, which in turn broadens the stopband bandwidths. The stopband bandwidth increases with increasing values of Z0e and Z0o, resulting in a compact E-PCML with smaller line widths and spacing. As an example, with Z0e/Z0o=2, the computed DBBSF responses for three different combinations of Z0e and Z0o are shown in Fig. 3. The variation of the first stopband 3 dB fractional bandwidth (FBW) with E-PCML impedances is shown in Fig. 4. These curves show that a wide bandwidth (>50%) is achievable. The FBW increases with increasing impedance values of the E-PCML. For all cases in Figs. 3 and 4, fC=1 GHz, and ZR=2 (Z1=50 Ω, Z2=100 Ω). The

FBW can be increased further by decreasing impedance values of OSIR for any fixed ZR, Z0e, and Z0o.

III. Fabrication and Measurements

Following the guidelines described in the previous section, a compact DBBSF operating at 0.9 GHz and 2.1 GHz with 3 dB FBWs of 72% and 36%, respectively, is fabricated on a low cost FR4 substrate with a thickness of 1.58 mm, εr = 4.3, and

150 Vamsi Krishnaet al. ETRI Journal, Volume 32, Number 1, February 2010 Fig. 5. Photograph of the fabricated filter.

Unit: mm 28.94

29.32 0.5

0.42

0.35 2.0

0.54 1.58

I/O feed lines 2.24

Table 1. Comparison with the best reported dualband BSFs.

Reference Frequency (GHz) Normalized circuit size

[1] Multiple bands Not available

[2] Multiple bands Not available

[3] f1=1.7, f2=2.30, fC= 4.0 0.0474 × 0.0411 [4] f1=1.5, f2=3.15, fC= 2.325 0.0056 × 0.0047 This work f1=0.9, f2=2.10, fC= 1.5 0.0024 × 0.0024

*Note: Normalized circuit size = (length/λg) × (width/λg)

tanδ = 0.022. All electrical lengths (θ1, θ2, and θ) are π/2 at fC=1.5 GHz. The impedance values are Z2=120 Ω, Z1=62 Ω, Z0e=147 Ω, and Z_0o=70 Ω. A photograph of the fabricated filter is shown with its dimensions in Fig. 5. The compact filter occupies an area of 28.94 mm × 29.32 mm. For comparison, the normalized circuit sizes of the proposed and the published filters are shown in Table 1, where λg is the guided wavelength at fC.

A full-wave simulator IE3D is used for EM simulations.

Measurements were carried out using the Agilent 8510C vector network analyzer. The measured, full-wave simulated, and computed responses of the proposed DBBSF are shown in Fig. 6. The E-PCML is assumed to have equal phase velocities in even and odd modes for which the approximation θe=θo is achieved by cutting a rectangular groove [5] 0.35 mm×2.0 mm on each line of the EPCML section. The measured stopbands with rejections of more than 35 dB are obtained at f1=0.92 GHz and f2=2.16 GHz with 3 dB FBWs of 68.6% and 36.1%. The measured insertion loss (IL), including the connector loss, is within 0.5 dB from DC to 0.44 GHz in the lower passband.

The 3 dB passband between the two stopbands extends from1.18 GHz to 1.75 GHz, where the IL is within 1.5 dB from 1.23 GHz to 1.56 GHz. The present FR4 substrate is lossy.

The substrate loss increases with increasing frequency. The IL

Fig. 6. Measured, full-wave simulated, and circuit predicted responses of the proposed DBBSF.

0 0.5 1.0 1.5 2.0 2.5 3.0 3.5

-50 -40 -30 -20 -10 0

Frequency (GHz)

S-parameters (dB)

S21

S11

Measured Fullwave Circuit

is slightly high in the upper passband and is within 2.5 dB from 2.55 GHz to 3.44 GHz.

IV. Conclusion

A compact DBBSF was developed using an end-shorted parallel-coupled microstrip line and open-ended stepped impedance resonators. Design guidelines were provided. The stopband frequencies and bandwidths are controlled by the impedances. A compact geometry was chosen so that the filter would occupy a rectangular area of 0.247 λg × 0.250 λg, where λg is the microstrip line guided wavelength at fC = 1.5 GHz. As the design uses only transmission line sections, the structure is simple and easy to fabricate, and the geometry is more compact than those of the filters reported in [1]-[4].

References

[1] H. Uchida et al., “Dual-Band-Rejection Filter for Distortion Reduction in RF Transmitters,” IEEE Trans. Microw. Theory Tech., vol. 52, no. 11, Nov. 2004, pp. 2550-2556.

[2] K. Rambabu et al., “Design of Multiple-Stopband Filters for Interference Suppression in UWB Applications,” IEEE Trans.

Microw. Theory Tech., vol. 54, no. 8, Aug. 2006, pp. 3333-3338.

[3] Z. Ma et al., “Novel Microstrip Dual-Band Bandstop Filter with Controllable Dual-Stopband Response,” Proc. Asia-Pacific Microw. Conf., Dec. 2007, pp. 1177-1180.

[4] K.S. Chin, J.H. Yeh, and S.H. Chao, “Compact Dual-Band Bandstop Filters Using Stepped-Impedance Resonators,” IEEE Microw.

Wireless Compon., Lett., vol. 17, no. 12, Dec. 2007, pp. 849-851.

[5] M.K. Mandal, K. Divyabramham, and S. Sanyal, “Compact Wideband Bandstop Filters with Sharp Rejection Characteristic,”

IEEE Microw. Wireless Compon., Lett., vol. 18, no. 10, Oct. 2008, pp. 665-667.