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SOFT SWITCHING FOUR QUADRANT DC-LINK INVERTER GENERATING SINUSOIDAL WAVEFORMS

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SOFT SWITCHING FOUR QUADRANT DC-LINK INVERTER GENERATING SINUSOIDAL WAVEFORMS

Byeong 0. Woo GoldStar Industrial System Co. Ltd.

533 Hogye-dong, Anyang-shi Kyongki-do 430-080, KOREA (FAX : 82-343-53-6263)

ABSTRACT

A new so t switching converter topology for high frequency ACI

6

CIAC power conversion is proposed, which synthesizes the sinusoidal output volta4es and results in the dramatic reduction o acoustic noise. The new converter achieves sop switc

R

ing without increas- ing voltage or current ratings of devices and has capability of high frequency and four quadrant opera- tion. The high frequency operation by soft switching provides several advantages such as high power density, low EMI and high dynamic perJ4ormance. The voltage stresses and the current stresses in this con- verter are limited to the supply voltage and the link current respectively by the o eration of current free- wheeling circuit, and thus t& V A ratings of devices are reduced to those of the hard switched inverters.

Operational principles, analyses and control method of the proposed converter are presented. Simulation results are also shown to verifr the operational principle.

INTRODUCTION

Man soft switching converters have received considerabL attention to replace the conventional hard-switched converters. The increase of a switching frequency resulted from eliminating switching losses by soft switching shows distinctive advantages such as high power densit , low EMI, low acoustic noise and high dynamic pedormance. Resonant circuit adopted to achieve soft switching unfortunately always invokes substantial penalties such as higher VA ratings for devices and handing a multiple of the load power due to an increase of current and/or voltage. It has ham- pered high power converters with soft switchin although many converter topolo ies are quite wefi suited to the ower conversion in L e low power range to be successPu1 ones.

A lot of converter topologies[l-141 have been suggested and studied to apply soft switching tech- nique to the high power conversion. The resonant pole inverter has the advantage of inte rating the resonant component and the output filters. however the power range is limited due to high current stress and losses in devices and components. The auxiliary resonant com- mutated pole inverter[4] has the attractive features of soft switching without additional voltage and current stress in devices and low losses in additional passive

'

Gyu H. Cho

Dept. of Electrical Eng., KAIST 373-1 Kusong-dong, Yusong-gu

Taejon 305-701, KOREA (FAX : 82-42-829-3410)

elements. Unfortunately it requires twice device counts to be not cost effective compared to hard switched inverter. The actively clam ed resonant dc-link invert- er[2] re uiring only one aiditional devices and limit- ing the (!levice voltage stresses to 1.3-1.5Vs has been considered to be notable renewal in high power soft switching approach. However it has a harmonic com- gonent be!ow .the switching frequency and thus needs a igher switching frequency to obtain the same perfor- mance as the conventional PWM inverter because of a discrete pulse modulation.

An new converter topology which reduces the VA ratings of devices and has excellent output wave- forms by inherent output filter capacitor was ro- posed[ l]..The idea of roviding a current freewheeying path in circuits for soyt switchin in order to limit an excessive current or voltage builf-up is attractive. The insertion of current freewheeling circuit makes it possible to separate the main ower flow interval from the resonant operation so Slat the VA ratings of devices are remarkably reduced. However the mode transition between the powerin operation and the regenerative operation requires a%ditional two switch- es except freewheeling switch in the link and it complicates the power circuits and converter control.

converter generatin sinusoidal output voltages witf only one additionaf switch is proposed which preserves the desired features of the previously pro osed-convert- er[l]. The existence of a common m o 8 between the powering operation and the regenerative operation simplifies the ower circuit and alleviates the control complexity. d e voltage and current stresses of de- vices are limited to the supply voltage and the link current respectively b the operation of a current freewheeling circuit. d i s converter is capable of four quadrant operation and dramatically reduces acoustic

In this paper a new soft switchin

I I T T

m

I I l l

Fig. 1. Proposed converter configuration

(2)

-6

Fig. 2. Equivalent circuit

noise due to the sinusoidal output volta e. Operational ed in letail in this paper.

princi les, analyses and simulation resu

f

ts are present-

PROPOSED CONVERTER

The proposed converter is shown in Fig. 1. The front-end converter is controlled for its output to be one of among zero, maximum(Ed) and minimum(-E,) of line-to-line supply voltage by selecting the switches proper1 while maintaining the input power factor near unity. h e inductor L, is much larger than L, to be approximated by a current source and the output filter capacitor CO is also much larger than C, to be approxi- mated by a voltage source during one switching c cle.

Since the main current flow on the link is a L a y s positive direction from the input to the output, turn- pg-on of the upper switch among one of output inverter legs causes the corresponding phase voltage to increase, on the other hand, turning-on of the low art switch causes the phase voltage to decrease. PereFore it is possible to synthesize the wanted sinusoidal output voltage by roperly selecting the inverter switch on eve switc\in cycle. The operation modes direct1 depen? on the pof arity of the reflected voltage from &e output capacitor to the link and are divided into owering operation(V,>O) and regenerative one(V, CO). %he power flow is forward from the inputaline to the output capacitor during the powering operation and backward durin the regenerative operation. Fig.2 shows the equiv 8ent circuit of the proposed converter.

PRINCIPLE OF OPERATION

Fi

.

3 shows the switch status and waveforms during &e powenn operation that consists of six modes, where the v8tage increment of output capaci- tor during one switching cycle is neglected for the simplicity of explanation. Assume that the current of inductor L is I&) and freewheels through SF and ca acitor

6,

is changed to ( E d

-

V,) while the current of iniuctor L, is zero.

1) Mode P1 .(rl ,tz> : The inductor curfent I&) increases rapid1 since the freewheelin switch con- ducts and the wxole voltage across the i n k is a plied to the inductor L,. During this interval the inxuctor current and the capacitor voltage are given by

=I&) (1.a)

s1

s2

I Lf

SF

I !i

I I . .

::*+

- . . ~

!: vo

.I

I: . .

.

l!!

iI I 1 I !

1 1 I 1 I

P1 P2 P3 P4 P5 P6

I

-T-

the equivalent circuit during powering operation Fig. 3. Switch statuses and voltage and current waveforms for

(1.b) L,

Vcr(r) = Ed - vo (1.c)

After a while the resonant inductor current 14,(t) is limited to the link current I&) and the freewheeling switch SF is turned off under zero current.

2) Mode P2 ( h , t3) : Since the switch SF is turned off, the resonant inductor L, and the link inductor L, effective1 .work in series. The current of series connected( inductor increase slowly because of large inductance.

In this interval

IL(t)=-(Ed-V,)f+I'i(fl) 1

I',(t) = Iy(f) (2.6)

V,, = Ed - V, (2.c)

Neglecting the voltage increment the equation (2.a) represents that the inductor current increase linearly during this interval. When the inductor current reaches the wanted value IF(k

+

1)

+

Al, this mode ends by turning-off the switch S,.

(t3,r,) : The capacitor C, is almost linear1 d r g e d s m e this interval is short enough to n e g i c t .the inductor current variation.

The equation are given by

(3.a 1

I'm

= I&) (3.b 1

(3.c) Vc,(t) E --(IF&)

+

N)t + E d -

v,

3) Mod

I&) s I&) +I&) = N

1

c,

(3)

s1

SZ I Lf

SF

111

Vc,

Vl

v2 mode

I

I

. . .

1 1 I 1 1 1 I @ 1 1 I 1

I ( I I I

I . . .

R1 R2 R3 R4 R5 R6

- T-

Fig. 4. Switch statuses and voltage and current waveforms for the equivalent circuit during regenerative operation

As soon as capacitor voltage reaches -V, the front-end switch S, conducts under zero voltage through point B.

4) Mode p4 (r4,r5) : The current of series connected inductor decreases almost linearl. since the output capacitor voltage is reversely appled to the inductor in reverse direction.

Thus

I',(f) =I&) (4.b 1

Vc,(r) =

-v,

(4.c)

This mode ends by turning-on S, when I&) is decreased to the wanted value I& + 1) that is the freewheeling current during next commutation inter- val.

5 ) Mode P5 ( r 5 , t 6 ) : In this interval the inductor current I&) starts to freewheel through S,. The current I,&) decays abruptly since output capacitor voltage is reversely applied to inductor L,.

The equations are given by.

I& = I,(k

+

1) (5.a)

I L , ( t ) = - - V J +I',OJ (5.b)

Vc,(r) =

-v,

(5.c)

This mode ends when the inductor current

1 L,

is reduced to IcoM defined as following;

where Z,+

6 ) Mode

N

(r,, r7) : Inductor current I t ) contin- ue to freewheel and the trapped energy to infuctor L, is transferred the capacitor by resonance.

I&) =I,&

+

1) (6.a )

I&) = I, sin o,r +IL,(r6) cos o,r (6.6 1

Vc,(t) =r,,(16coso,t 1 -ILr(t6)sin a,t) ( 6 . ~ 1 where Vo

5

When I ( t ) becomes zero, reaches its peak value.

From the energy conseryation and (5.d), the peak value is given by

I - _ -

6 -

= Ed

the voltage Vc,(r) equation (5.c) and

After mode 6 ends, all states are the same as those of the beginning of model. Therefore, a next switching cycle can start again.

As mentioned above the powerin and the regenerative operation are classified accorjin to the direction of the power flow between the input fine and output capacitors. In the re enerative operation the power flow is reversed from tfe output capacitor to the in ut line. The regenerative operation is similar to that o f t h e powering operation except that the e uivalent front-end converter switch is switched to (B(durin model and mode2) and then A(during mode3 a n i mode4) in sequence. During the regenerative operation the switch status and waveforms are shown in Fig. 4 and analytical expressions are summarized in Ap- pendix.

CHARACTERISTIC

The volta e stresses of the devices and compo- nents are limitet to the maximum supply voltage and the current stress is always clamped to the link current.

Therefore the VA ratings of this converter are ver low compared with those of any other resonant l i d inverters. Sinusoidal voltages for both sides rovide high quality waveforms to. the input and to t i e load and result in lower harmonic current and lower acous- tic noise in the motor drive system.

LINK CURRENT CONTROL

From the equation (2.a) and (4.a), the link current vanes dunng mode2 and mode4 as followings, ( 5 . 4

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A l a

I

At 4 6 I I

-

I I

4’

*

t Fig. 5 . Link current variation during one switching cycle

@ @ @

@-GI

I

b-6

Powering

i

Regenerative

mode I mode

!

Fig. 6. Mode transition diagram

Since mode3 is short enough to ne lect the current variation and the current freewheek during model, mode5 and mode6, the link current variation for one switching cycle is given by,

Where I,(&) and I,(& + 1) represents the free- wheeling current in k’th and (k+l)’th switching cycle.

Fig.5 shows the current control method by the ratio of Arz and Ar4 under a constant switching cycle.

THREE PHASE CONFIGURATION A. Mode Transition

The proposed converter synthesizes sinusoidal voltages at the output capacitor by impressing the high frequency link current c cle by cycle to the selected output capacitor. It shouh be noted that the selection of the inverter switches is done by observing the errors. Therefore, the reflected voltage(V,) from the output capacitor to the link is vaned cycle by cycle and its polarity also can be changed. Considering that the operation mode of powering and regenerative opera- tion are directly de endent on the voltage olarity of an inverter input voytage, it is necessary to {ave mode transition capability between the .powering and regen- erative operations. Fortunately in this converter the

Fig. 7. Three phase configuration for output capacitor bank and load where phase A and B are compensated

v

t k

4

- 7

k + l t

I

I I

t

‘Lf

freewheeling I I I 1

1-

, I

I i l

t Fig. 8. Compensation process for phase A and B

powering and regenerative o eration have a common mode(mode 6 ) as shown in Ag. 6. Thus it do not need any other action for a mode transitjon. By delivering or extraqing the power in each individual high frequency cycle it is possible to synthesize a sinusoidal output voltage independently to the load condition.

(5)

compensated Va

---______

I

uncompensated /

I I I

t k+l t k.

Fig. 9. Compensation process based on one step ahead errors calculation

--

B. Error Compensation with Present Data

To make an output capacitor voltage follow the sinusoidal references, phase switches conducting dur- in next switching cyc e should be determined at mode 6 . h r e e errors between the reference voltage and the capacitor voltages are calculated.

Thus

v.'

-

v,

= E,

vc*

-

v,

= E,

and

Ea + E, + E, = 0

The selection principle of phases compensated during next switching cycle are as followings : the hase having the largest error out of three phase is Firstly chosen and then the phase having the lar er error amon the opposite polarity error i s selected. &e upper swift% corresponding to a positive error phase and the lower switch of the phase with a negative error conduct to compensate both hase errors. Fig. 7 shows three phase configuration r f! output capacitor bank and load where phase A and B are compensated.

Compensation process for phase A are shown in Fig.

8.

--

C. Error ComDensation y& Prediction

By measuring the load currents together with output voltages, it is sufficient to know the uncompen- sated output voltage without of the next state,

1

V'Jfk+l) =

va(fk)+-/07-

CO

(12)

Above equations show the calculation of a redicted output voltage for the next state. Since we L o w the estimated output voltage and the reference voltages, we can obtain the expected error quantities

for the next state before the next switching occurs,

' . ' ( l k + I ) - v a ( f k ) = E:

Based on one ste ahead errors instead of present values, we can a g o determine which phases should be chosen for the next state. Fi

.

9 shows three phase reference voltages, the expectef output voltage without compensation and the actual output voltages with com ensation based on one ahead errors. It can be seen $at the selection of phases compensated is based on error values at not f = r k but. t = f k + l . From simulations better results were obtained in this way.

SIMULATION RESULTS

The pro osed converter is simulated to verify the operation. k p u t AC source. voltage is 300V,,, and the parameters used for simulations are as followings ;

L, = 30w Lf = ImH C, = 30nF CO = 3 0 p

Simulation waveforms obtained with this con- verter are shown in Fig. 10. The simulation results are in good agreement with the expected waveforms as shown in Fig. 3 and Fig. 4. Fig. 11 shows the simulation results for the inverter output volta e waveforms when the command is abru tly changed.Tt can be seen that transient interval is sgort and a very fast response is obtained. As expected previously the simulation results show that the voltage stresses on all of the devices are almost e ual to source voltage and the peak current stresses areqimited to the link current.

CONCLUSION

A new high fre uency dc-link converter gener- ating sinusoidal wave2orms for both input and output sides with soft switching capability is proposed and verified by the simulation. The proposed converter can operate on full four quadrants and provides several improved characteristics over the other resonant link inverter such as low VA ratin s, low acoustic noise and low harmonic distortion. d e volta e and current stresses are also low and always limitei to the maxi- mum of supply voltage and the link current respective- ly. The front-end converter can operate on almost unity power factor as well.

The features of the proposed converter can be summa- rized as follows :

*

elimination of switching losses due to soft

*

low VA ratings of devices

*

four quadrant operation

*

low harmonic on both input and output side switching

(6)

*

capability of constant frequency operation

*

simple power structure

*

fast response and low acoustic noise

*

four uadrant o eration

*

suitabPe for hi& power level

A X )

1 5 - 1 , : , . , , I

l S < ' ! ,

-

1Jt)o. t. * t , i 4

:

3 , -

I , , , a m -

, ,

2 0 -

10- i , , . , , , , . -

5 . , / , , I , . : I -

3

- 5 - ! . , . , , a 8 % -

-10- 8 , . I #

- 1 5 - t I t , I /

3 -200

2

[msecl 0.k 0.h 0.k 0 08 0:I 0 . h 0.;4 0,;s 0.;8 0 2

V

"G

[msecl (d) output voltage

Fig. 10. Simulated waveforms

300 200 100

-100 -200 -300

4ww

-500 0 2 4 6 8 10 12 14 16 18 20

[msecl Fig. 11. Reference and actual output voltage waveforms for the

change of output voltage references (b) resonant inductor current

REFERENCES

400

I

400A

0.02 0.04 0.06 0.08 0:l 0.;2 0.;4 0:lS 0:lS 6.2

[msec]

(c) resonant capacitor voltage

[1] B. 0. Woo and G. H. Cho, "Soft switching AC/DC/AC converter with current freewheeling cir- cuit", IEEE PESC Rec., p. 31-38, 1991.

21

D.

M. Divan and

8.

Skibinski, '.'Zefo switchin loss inverters for hi h power applications", IEEE Trans. Ind. Appl., vof25, no.4, pp. 634-643,JulylAu-

ust. 1989.

f3] Y. Murai and T. A. Lipo, "High fre uency series resonant dc link Dower conversion", IE& IAS Rec.,

.772-779, 198g.

r!]

R. W. De Doncker and J. P. L ons, "The auxiliary resonant commutated converter",

f

EEE IAS Rec., pp. ..

1228-1235,1990.

[ 5 ] Y. Murai et al, "Current pulse control of high fre uenc series resonant dc link ower converter", IE& IA8 Rec., pp. 1021-1036,1988

(7)

[6] Jih-Sheng Lai and B. K. Bose, "High fre uency quasi-resonant dc volta e notchin inverter ?or ac motor drives", IEEE IASSec., pp. 18203-1217, 1990.

[7] B. 0. Woo, I. D. Kim and G. H . Cho, "Zero volta e switching AC/DC/AC converter using modi- fied Ri h fre uency dc-link", IEEE IAS Rec., pp.

1243-1550, 1980

[8] S . S. Park and G. H. Cho, "A current regulated pulse width modulation method with new series reso- nant inverter", IEEE IAS Rec., pp. 1045-1051, 1989.

[9] E. C. Nho and G . H. Cho, "A new high perfor- mance zero-voltage zero-current mixed mode switch D C D C converter , IEEE PESC, p. 902-908,1989.

[IO] I. D. Kim and G. H. d o , "New bilateral zero-voltage switching AC/AC converter usin high frequenc artial resonant link", IEEE IECOPf Rec., ffl] D. M. Divan, "Resonant dc link converter - a new conce t in static ower conversion", IEEE IAS Rec., ff2] G. Venkataramknan and D. M. Divan, "Pulse width rnodulatlon with resonant dc link converters", IEEE IAS Rec .984-990, 1990.

[13]

D.

I. A."k%tens and D. M. Divan. "A hieh .857-{66p, 1990.

. 6 & - 6 5 6 , 1 9 8 8

fre uency resonant dc link inverter using. IGBT'?, IEZof Ja an IPEC Rec., p

.

152-160, 1990.

[14] S.

2

SUI and T. A. fipo, "Field oriented control of an induction machine in a high frequency link ower system", IEEE PESC Rec., pp. 1084-1090, P988.

APPENDIX

Analytical expressions for the link current, the resonant inductor current and resonant capacitor volt- age of the six modes during regenerative operation are given as follows. (V,<O)

1 ) Mode R 1 : (rl , r2)

I,(! 1 = I , (k 1 ( A . l a )

( A . l b ) ( A . l c )

VO L/ + L,

fdr

= - t + I # ) (A . 2 a )

IL,(-t) =I,&) (A .2b)

(A .2c)

(A . 3 a ) (A . 3 b )

4) Mode R4 : (rs , rJ

JL,(f) =

-- (Ed+v4

+I/(k

+

1)

L, Vc,(f) = -Ed - V , 6) Mode R6 : ( r 6 , r,)

I&) =I#

+

1)

(A .4a ) (A .4b ) (A .4c)

(A .5a)

(A .5b) (A .5c)

(A .6a)

(A .6b ) (A .6c)

(A . 3 c )

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